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# Theory of Linear and nonlinear circuits - Engberg J.

Engberg J. Theory of Linear and nonlinear circuits - Wiley & sons , 1995. - 154 p.
ISBN 0-47-94825 Previous << 1 .. 26 27 28 29 30 31 < 32 > 33 34 35 36 37 38 .. 85 >> Next The formulae, which are derived in [i], are given below. The unprimed entities are the small signal and noise pnrnmetprs of the three-port without new embedding, and the primed entities are the parameters with a set oi embedding demerits.
Parallel embedding:
Y,\ = Yu + Ya 4- YB
(6.1)
6.1. Lumped embedding
10.3
(6.3)
ąŻ22 + ąŻ ą▓ + ąŻčü ąĢ,,
ąÜ
(6.5)
(6.6)
ąæ, DVE,
(6.7)
where
Dy = |ąŻąÆ - ąŻąĮ!2
By = Gb + Gc + 1^21 |2ąöą┐
čÅ, = (ąŻčå + ąŻ-i + Yi\)Gb + (ąŻą┐ + ąŻ 4 + Ya)Gc
+ (ąŻčå ŌĆö ąŻ ąÆ ^21 Rn + (ąŻ-t + ąŻ ąÆ + V-,; ąŻ ą╝ ! ŌĆś 1(ą┐
Ly = | i's ŌĆö ąŻ'21I*" (G ą░ + Gn) -f- |ąŻą┐ f ąŻ4 + Y2\V Gb
+ |ąŻą┐ + ąŻ ą░ + ąŻą▓^ąĪčü + |(ąŻčå ŌĆö ąŻ-,) ąŻą▓ + (ąöą╗ + ąŻ ą▓ + ąŻ-^ąŻą╗!2 Rn
Series embedding:
(6.12)
(6.8)
(6.9)
16.10)
(6.11)
(6ąøąŚ)
Lz \H: \
Ds D, ┬ŻŌ¢Ā
┬Żą│ = Rb + Rc -f \^2\\~9n
H, ŌĆö (Z11 -f Za ŌĆö Z21) Rb + (2čå + Za x Zg)Rc
104
6. Noise of embedded networks
Lz ŌĆö !Zg + Z2\\2{Ra + rn) 4- |Zu f Za - Z2\\2 Rb + \Zn + ZA + Zb\2Rc +- i(Z~, - Zu)Zb + (Za + Zb + Z,t) Z2^\ą│ą┤ą┐
Transformation from Z to Y parameters:
(6.15)
(6.16) (6.17)
Yn = Zn
ą¤ 2 = ŌĆö Z\2
Y21 = -Z21
Y22 = Zn
Rn = rn + gn\Z^\2
Gn = rn
\z-,\2 + rn!gv.
y7 = z;
|z7!2 + Tnl3n
where = Z\\Z22 ~~ ^12^21
Transformation from Y to Z parameters:
IKr + Orj'Rr,
ą¤ / R
(6.18)
(6.19)
(6.20) (6.21)
Zn = ~ (6.22)
ąöčā
Z12 = (6.23)
Zn = -r^- (6.24)
LAY
Z22 = ~ (6.25)
A čā
gn = Gn + flniv;!2 (6.26)
(6.27)
V'-
It is interesting to note that the formulae work both ways: embedding and deembeddins. As an example take an encapsulated transistor whose small signal
6.1. Lumped embedding
105
parameters and noise parameters are measured. The effect of (the known) strav capacitors and stray inductors can be removed by adding the negative of the strav elements. This is done such that the stray elements closest to the terminals of the encapsulated transistor are removed first and then the second set of stray elements until finally the small signal and noise parameters of the transistor chip are obtained.
Transistor
Ō¢Ā47/IS ^ 0.5 mS 30 fi
Figure 6.2: Transistor with bias and feedback and equivalent circuit. Capacitors are considered as shorts at the working frequency.
Example 6.1 A transistor amplifier stage as seen in Figure 6.2 consists ąŠčé a transistor with known Y and noise parameters and also bias and load resistors and finally a series feedback resistor. The data for the transistor are
I'll = 10 4- j 2.1 mS
Vis = 0.5 ŌĆö j 0.866 mS
Yii = 125 - j 29 mS
y22 = 1 + i 3 mS ,J r-
R., = 25 0 = 4.'? mS
y. - ŌĆś> -4- 7 7 5 mS
These data give Fmin - 1.3 and i 'sof ŌĆö 14 ŌĆö j 7.5mS. in order to compute the small signal and noise parameters of the stage, one has to apply Equations (6.22) - (6.28) to get the transistor data into Z form. Then Equations (6.8) - (6.14) add the series feedback resistor (whiie Z4 = Zc = 0 + jO). Equations (6.15)- (6.21) transform to
106
6. Noise of embedded networks
the Y parameters and Equations (6.1) - (6.7) add the bias and collector resistor (with Yb ŌĆö 0). The final result, computed with a Fortran program where the equations used are shown as subroutines in Appendix D, is:
Yn = 1.586 4- j 1.548 mS
Yu = 0.4448 - j 1.017 mS
Yu = 25.16 - j 2.935 mS
- -* ex. 0.2837 -{- j 1.469 ŌĆö ąĪ .UlU
Rn = 62.81 0
G'ŌĆ× = 4.735 mS
K, = ŌĆó3.442 + j 3.432 mS
= 2.606 and Ysof = 9.340 ŌĆö 3.432 mS. The main part
of this noise degregation is due to the series feedback. The bias and collector resistors raise the noise factor only slightly - from 1.800 to 1.807.
Figure 6.3: Feedback elements for simultaneous input power match and noise optimization.
Example 6.2 An optimization procedure for simultaneous input power match and noise optimization using only one series and one shunt feedback element as shown in Figure 6.3 has been developed. Since lossless feedback does not change the value of the minimum noise measure (does not add noise), only lossless elements are considered. The fact that ą£,.čéą½ is constant for lossless feedback is a convenient check on results of computer programs. What dees change, of course, is the value of the source admittance, YsoM: minimum noise measure.
For each pair of iossiess feedback elements a computer program - using the subroutines in Appendix D - gives the values for the primed signal and noise parameters from Equations (6.1) - (6.28). The minimum noise measure, ą£ąĄčéčŗ, and the corresponding source admittance, YsoM- from Equations (5.27), (5.?5) and (5.26) are also computed.
6.1. Lumped embedding
107
-1 0 1
ImfV'/l mS
Figure 6.4: Transducer gain Gt (full lines) and load admittance Gl, Bl (dotted lines) versus lossless feedback for unity VSWR and minimum noise measure.
When the source admittance, Ys = Ysom <s determined, input power matching occurs when the load admittance, Yl, 's chosen to satisfy the relation:
>L - ŌĆö>22 + -----------777
ąŻąĮ ŌĆö > čÅ
This equation allows choosing the load admittance for input power match and minimum noise measure (V^oa/). Then, with the source and load admittances equal to Yso\f and YiQ\f respectively, the transducer gain, Cj, of the stage is computed. Previous << 1 .. 26 27 28 29 30 31 < 32 > 33 34 35 36 37 38 .. 85 >> Next 